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Optimal bounded-influence tests can be obtained by extending Hampel's optimality criterion for estimators (see Section 12.2) by finding a test in a given class that maximizes the asymptotic power at the model, subject to a bound on the level and power influence functions. If the test statistic T is Fisher-consistent, that is, ('(00) = 1, then E-l = V(F0,. T ) , the asymptotic variance of the test statistic. Thus, finding the test that maximizes the asymptotic power at the model, subject to a bound on the level and power influence function, is equivalent to finding an estimator T that minimizes the asymptotic variance, subject to a bound on the absolute value of its self-standardized influence function. The class of solutions for different bounds is the same for all levels, and it does not depend on the distance of the alternative A. Therefore, the optimal bounded-influence test is Uniformly Most Powerful. A similar



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The peak inrush current the ( n + 1)

[l41 N. Anderson and P. Howard, Technology and Transceiver Architecture Considerations for Adaptive Antenna Systems, in Proceedings of ACTS Summit, pp. 965-970, 1997. [l51 J. Strandell, M. Wennstrom, A. Rydberg, T. Oberg, 0. Gladh, L. Rexberg, E. Sandberg, B. V. Anderson, and M. Appelgren, Experimental Evaluation of an Adaptive Antenna for a TDMA Mobile Telephony System, inProceedings of PIMRC, pp. 7984, 1997. 1161 J. J. Monot, J. Thibault, P. Chevalier, F. Pipon, S. Mayrargue, and A. Levy, A fully programmable prototype for the experimentation of the SDMA concept and and use of smart antennas for UMTS and GSMDCS 1800 networks, in Proceedings of PIMRC, (Helsinki, Finland), pp. 534-538, September 1997.





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The noise gure of a (ideal) passive attenuator at a temperature of T0 (290 K) equals its attenuation. This is because the available noise at the output of the attenuator is the available noise from the Thevenin resistance of the attenuator, presumably the same as the standard impedance of the cables at that point in the cascade. This is the same as the available noise from the source, at the input to the attenuator, during characterization. Thus the noises in Eq. (3.1) cancel and f becomes the ratio of input signal power to output signal power, which equals the attenuation. If we did a circuit-noise analysis of an attenuator, say a or T network, we would get the same results (but less ef ciently). We can do it either way (but must not add the two effects). The combined noise gure of a module preceded by an attenuator at T0 equals the module noise gure plus the attenuation. (The gain of the combination is, of course, lowered by the attenuation also.) To see this, write Eq. (3.14) for an attenuator followed by a module, using 1/g1 for the attenuation of the attenuator: f = f2 1 f2 1 + = . g1 g1 g1 (3.24)

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SPLIT-FERRITE CURRENT PROBE* Split-ferrite cylinders (or clamp-on ferrite beads) can be used as a calibrated current probe when clamped over a cable of interest. Common-mode currents are useful for measuring both frequency and amplitude components that may cause radiated EMI to exist above acceptable levels. This probe produces a voltage output that is related

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Now another relation to achieve maximum power transportation from the source to the load resistor can be found from the equation (9.62). Mathematically, by partially differentiating (9.62) in respect to RL, RL PRL 1 2 2 R RL = vS = vS S 2 . 2 3 RL ( RS + RL ) ( RS + RL )3 ( RS + RL ) (9.63)

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Here Eqs. (3) and (5) have been used. From this we obtain the actual power gain (Appendix G) for a cascade of N modules:

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